Communications radios, such as those employed in mobile cellular handsets and similar devices, generally consist of a transmitter and receiver. The transmitter and receiver are active simultaneously when a full-duplex communications standard is employed. The transmitter in a mobile radio device is generally power-controlled, so that the resulting power in the transmitted signal is just sufficient to be received and decoded at the basestation. In addition, modern communications standards typically employ modulations with a substantial variation in transmitted amplitude from one symbol to the next: that is, the peak-to-average power ratio (also known as the crest factor) is much greater than 1. When transmission occurs at high average power with high peak-to-average ratio, substantial advantages in overall efficiency can be obtained by varying the voltage supplied to the power amplifier in synchrony with the amplitude of the transmitted signal; this approach is generally known as Envelope Tracking (ET) when the power amplifier is operating in a substantially linear regime, or Polar Modulation when the power amplifier is fully saturated.
Envelope Tracking operation requires that the power supply for the RF power amplifier adapt its output rapidly. For example, in WCDMA operation, a chip lasts approximately 260 nsec, so a new optimal output voltage may be required as often as four times per microsecond. Even higher rates may be required in the Long-Term Evolution (LTE) standard. Envelope Tracking power supplies are typically constructed using either a linear amplifier, a hybrid combination of a slow switched power supply and a fast linear amplifier, or a switched power supply with very high switching frequency and control bandwidth.
A fast switched power supply will produce noise at the switching frequency and its harmonics; any phase noise in the switching frequency is multiplied by the order of the harmonic. In addition, for a switched ET supply, envelope modulation of the PWM (pulse-width modulated) signal results in spectral spreading of the switching frequency and its harmonics, as described, for example, by du Toit Mouton and Putzeys. The fundamental is (roughly) spread by the envelope bandwidth, and spreading at the higher harmonics increases linearly with the harmonic multiple n. Finally, some broadband thermal and (1/f) noise is expected to be present, primarily from the switching edges, which being abrupt contain a broad discrete frequency spectrum, spread by any dither present in the switch timing. When the switches are fully on or fully off, little noise is generated.
The radio receiver must be capable of receiving very small signals with small relative bandwidth, and is thus sensitive to periodic disturbances of any kind whose frequency lies within the wanted received channel. In full-duplex operation, where a transmitter sends signals in one band at the same time that a receiver is trying to receive signals in a paired channel of a second band, the transmit and receive frequencies vary, but the frequency separation between the transmit and receive channels is generally a fixed value, fduplex=fRX−fTX. (Note that in most cases the mobile unit transmits in the lower and receives in the higher of the paired frequency bands. In a few cases, such as bands 13, 14, and 20 of the 3GPP Technical Specification 36.101, the receive band (channel) is lower than the transmit band (channel). Since most nonlinear mixing processes produce roughly equal mixing products at the sum and difference of the carrier and modulating frequency, the discussion below generally applies to either approach to allocating transmit and receive bands.) When transmission and reception are simultaneously occurring, signals present in the transmit chain may result in desensitization of the receiver. A spurious signal from the transmitter at the frequency the receiver is tuned to may impair the ability to recognize and decode the wanted signal. Extremely small spurious signals can cause problems. For example, consider a WCDMA receiver with sensitivity is limited by thermal noise. The signal bandwidth is about 3.8 MHz, so thermal noise entering the receiver at room temperature is about −174 dBm/Hz+66 dB≈−108 dBm. If the receiver noise figure is 8 dB, the noise floor in the receiver channel is about −100 dBm (that is, 10−13 Watts). Any spurious signal comparable to or larger than −100 dBm will degrade receiver sensitivity; the total power of added noise and spurious signals should be on the order of 10 dB below the receiver noise floor to achieve <1 dB of degradation in sensitivity. Since the transmitted signal may be as high as 30 dBm or higher, avoiding desensitization requires that noise resulting from the transmit chain be attenuated to −140 dB below the transmitted carrier (−140 dBc) at the receiver input.
Undesired spurious transmit signals can result when broadband noise undergoes nonlinear mixing with the carrier frequency, resulting in sum and difference frequencies which may overlap the paired receive channel. Sources of broadband noise, as noted above, include amplified thermal noise and switch clock jitter. FIG. 1 shows a transmitter power amplifier 115 and a Low-Noise receiver amplifier 135 of a transceiver, in which the transmitter power amplifier is powered by an envelope tracking power supply, and depicts the problem described above. Noise from an envelope tracking power supply 110, with a spectrum shown in inset 155, mixes with the intended output of the power amplifier 115 to produce an output spectrum containing both the wanted signal, and spurious signals at the sum and difference of the carrier and noise frequencies. An example portion of the spectrum showing spurious output at the sum of the carrier and noise frequencies is depicted in the inset 160. When the spurious output is displaced upwards from the carrier (FTX) by the duplex frequency (fduplex), as depicted in 160, the spurious frequency lies on top of the paired receive channel for the phone. This noise is partially rejected by the transmit band filter or duplexer, such as 120. Since the spurious frequency is within the intended receive band, it is not rejected by the receive band filter 130 and enters the Low-Noise Amplifier 135, where it may interfere with the wanted signal in the assigned receive channel. This interference results in desensitization of the Receiver 140 to the wanted signal. The transceiver further includes a transmitter 150, baseband circuitry 145, and an antenna 125. Further, a battery is connected to the envelope tracking power supply 110.
Noise that is not exactly at the duplex frequency may still lead to desensitization in the case where the bandwidth of the transmitted and received signals is a substantial fraction of the duplex separation. FIG. 2A shows a frequency domain that depicts an example, in which a receive channel is placed above a transmit channel by the duplex frequency fduplex. A transmitted signal 210 of bandwidth BW may mix with noise at frequency fnoise to create spurious output 230 over a range of frequencies BW or larger. If this spurious output overlaps the paired channel 240, desensitization may result even though fnoise is as much as BW less than fduplex. For example, the Frequency-Division-Duplex (FDD) version of the Long-Term Evolution (LTE) physical layer allows channel bandwidths, BW, of as much as 20 MHz.
Modern multicarrier communications systems may also allocate specific subcarrier regions (resource blocks) to specific uplink and downlink users. Thus, it is possible for a mobile radio to be assigned a transmit resource block 5 or even 10 MHz above the center of the transmit channel, and a receive resource block 5 or 10 MHz below the center of the receive channel. FIG. 2B shows another frequency domain that depicts an example wherein a resource block or blocks 250, centered above a nominal carrier 200, may be allocated to the mobile unit's transmitter, while a resource block or blocks 280 centered below the nominal duplex carrier may be allocated to the same mobile unit's receiver. Mixing with noise at frequency fnoise<fduplex then produces transmit spurious output 270, overlapping the wanted received signal in the allocated resource block 280. Noise entering the Power Amplifier at a frequency equal to (fduplex−fbb,TXoffset−fbb,RXoffset), where the quantities fbb,TXoffset and fbb,RXoffset are the offsets from the center of the channel to the assigned resource blocks, could then produce spurious output lying on the wanted received signal. The potential for desensitization is increased since the transmitted energy is concentrated in the relatively narrow allocated channel, increasing the net overlap between the spurious output and the wanted signal. This displacement can be particularly challenging when the resulting range of sensitive frequencies overlaps the nominal switching frequency, or a low harmonic thereof, for a switched ET supply.
In the case where the Envelope Tracking power supply is a switched power supply, the output of the switched regulator is filtered by series inductor and shunt capacitor. At relevant frequencies, a corresponding simplified schematic is depicted in FIG. 3A. A switched ET supply is represented by a simplified noise model 310. The voltage source Vnoise represents the high-frequency noise produced within the switched converter. An output inductor Lout and capacitor Cout act as a double-pole filter at low frequencies, but at the higher frequencies of interest for duplex noise, the equivalent series inductance ESL and resistance ESR of the capacitor and printed circuit board routing must be considered. The transmission 320 to the power amplifier (treated as a resistance of about 25 ohms for frequencies well above the envelope tracking bandwidth) is depicted in FIG. 3B, for representative component values for a 3-5 MHz bandwidth envelope tracking system, appropriate for e.g. WCDMA or low-bandwidth LTE transmission. Note that the inductor Lout includes a frequency-dependent loss term equivalent to about 0.15 ohms at low frequencies and 10 ohms at 100 MHz, representative of typical commercial surface-mount components. This loss term has modest effects on the predicted noise transfer function for the frequencies of interest. A maximum in transmission around 3 MHz results from the series resonance of Lout and Cout, which may be useful in adapting the overall response of the envelope tracking power supply, and must in any case be accounted for in design of the control system. A broad minimum in transmission results from the low-Q series resonance of Cout and ESL. For the value of ESR given in the figure, the minimum transmission is around −45 dB at 80 MHz, which may be insufficient to avoid receiver desensitization, depending on the emitted noise levels from the switched converter.
It is possible to insert an additional filter structure, such as a second series L-shunt C, a parallel L-C notch filter, or a more sophisticated multi-pole filter, between the envelope tracking power supply and the power amplifier branches. However, such a structures require additional components and board space. Each additional series inductance introduces additional losses, which degrade system efficiency, and each additional shunt capacitance increases losses in envelope tracking mode. The filter configuration must also ensure that the ET power supply presents a low impedance seen from the power amplifier, to ensure that current required by the varying RF output signal is available. In fixed-voltage configurations, a large decoupling capacitor, often several microFarads, is placed at each power amplifier supply input. As described in more detail below, such large capacitances result in unacceptable degradation of efficiency in envelope tracking operation.
It is therefore desirable to have methods and apparatuses for voltage regulation that provide both high efficiency envelope tracking and low paired-channel transmit noise, with minimal additional discrete components or printed circuit board space required.